Impulse noise reduction to an MCM signal

ABSTRACT

A method reduces noise in a multiple carrier modulated (MCM) signal. Such a method may include: estimating impulse noise based in the equalized signal; and removing a portion of the noise in the equalized signal as a function of the estimated impulse noise. An apparatus reduces noise in the MCM. Such an apparatus may include: a down-converter; an analog to digital converter to digitize the output of the down-converter; a guard-interval removing unit operable upon the digitized output of the down-converter; and a combined FFT, equalization and impulse-noise-compensation unit operable upon a signal from the guard-interval-removing unit.

BACKGROUND OF THE PRESENT INVENTION

Multiple carrier modulation (MCM) techniques, e.g., orthogonal frequencydivision multiplexing (OFDM), are generally known.

Like other communication techniques, a received MCM (again, e.g., OFDM)signal is subject to noise introduced between the transmitter and thereceiver. This noise includes additive white Gaussian noise (AWGN) andimpulse noise. Typically, non-negligible impulse noise sources areelectronic devices near the receiver that exhibit large local magneticfields induced by switching large amounts of current, e.g., duringenergization of: motors that move an elevator carriage; motors and/or anexposure device in a photocopier machine, etc.

It is known to estimate (and attempt to remove) impulse noise content inthe received multi-carrier-modulated (MCM) signal prior to the signalbeing equalized (a “pre-EQ signal). FIG. 1 is a block diagram depictinga typical MCM system 100 that includes components for removing impulsenoise from a pre-EQ signal, according to the Background Art. Forsimplicity, only a portion of the system 100 is depicted, as isindicated by the ellipses ( . . . ).

In more detail, system 100 includes the following serially-connectedcomponents: a down converter 126; a clipping unit 101; and aguard-interval removing unit 130. Clipping unit 101 includes: a variablegain amplifier 180; a clipping device 182; an analog-to-digitalconverter (ADC) 184; a feedback loop formed of a power estimation unit186 operating upon the output of ADC 184; and a threshold calculationunit 188 that operates upon the power estimate from unit 186 andprovides a threshold control signal to amplifier 180. Details concerningFIG. 1 can be found in published European patent application,publication No. EP 1011235. Magnitudes of the pre-EQ signal greater thanwhat is typically expected are detected and either clipped to athreshold level by clipping device 182, or to zero (e.g., publishedEuropean patent application, publication No. EP 1043874).

SUMMARY OF THE PRESENT INVENTION

At least one of the embodiments of the present invention is directed toa method of reducing noise in a multiple carrier modulated (MCM) signal.Such a method may include: estimating impulse noise based in theequalized signal; and removing a portion of the noise in the equalizedsignal as a function of the estimated impulse noise.

At least one other of the embodiments of the present invention isdirected to an apparatus for reducing noise in a multi-carrier-modulated(MCM) signal, the apparatus comprising: a down-converter; an analog todigital converter to digitize the output of the down-converter; aguard-interval removing unit operable upon the digitized output of thedown-converter; and a combined FFT, equalization andimpulse-noise-compensation unit operable upon a signal from theguard-interval-removing unit.

Additional features and advantages of the present invention will be morefully apparent from the following detailed description of exampleembodiments and the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram depicting a typical MCM system for pre-EQimpulse-noise removal, according to the Background Art.

The other drawings are: intended to depict example embodiments of thepresent invention and should not be interpreted to limit the scopethereof.

FIG. 2 is a block diagram depicting an MCM system according to at leastone embodiment of the present invention.

FIG. 3 is a more detailed block diagram of the combinedFFT_&_Equalization_&_Impulse-Noise-Compensation unit of FIG. 2,according to at least one embodiment of the present invention.

FIG. 4 is a more detailed block diagram of the peaks-detection unit ofFIG. 3, according to at least one embodiment of the present invention.

FIG. 5 is a block diagram of receiver, at a level of detail between thatof FIGS. 2 and 3, according to at least one embodiment of the presentinvention.

FIG. 6 is a block diagram of anFFT_&_Equalization_&_Impulse-Noise-Compensation unit 450, according toaccording to at least one embodiment of the present invention.

FIG. 7A is a block diagram of anFFT_&_Equalization_&_Impulse-Noise-Compensation unit according to atleast one embodiment of the present invention.

FIG. 7B is a block diagram visual-summary of a receiver, related to FIG.7A that corresponds to FIG. 5, according to at least one embodiment ofthe present invention.

FIG. 8A is a block diagram of aFFT_&_Equalization_&_Impulse-Noise-Compensation unit according to atleast one embodiment of the present invention.

FIG. 8B is a block diagram visual-summary of a receiver related to FIG.8A that corresponds to FIG. 7B.

FIG. 9A is a block diagram of aFFT_&_Equahization_&_Impulse-Noise-Compensation unit according to atleast one embodiment of the present invention.

FIG. 9B is a block diagram visual-summary of a receiver, related to FIG.9A that corresponds to FIG. 8B, according to at least one embodiment ofthe present invention.

FIG. 10 is a block diagram of anFFT_&_Equalization_&_Impulse-Noise-Compensation unit according toanother embodiment of the present invention.

FIG. 11A is a block diagram of aFFT_&_Equalization_&_Impulse-Noise-Compensation unit according to atleast one embodiment of the present invention.

FIG. 11B is a block diagram visual-summary of a receiver, related toFIG. 11A, according to at least one embodiment of the present invention.

FIG. 12A is a block diagram of aFFT_&_Equalization_&_Impulse-Noise-Compensation unit according to atleast one embodiment of the present invention.

FIG. 12B is a block diagram visual-summary of a receiver, related toFIG. 12A, according to at least one embodiment of the present invention.

DETAILED DESCRIPTION OF EXAMPLE EMBODIMENTS

In developing the present invention, the following problem with theBackground Art was recognized and the physics thereof was determined.Regarding pre-EQ noise-removal, much of the significant impulse noisecontent has magnitudes that are comparable to or less than the largesttypically-expected magnitudes in the pre-EQ received MCM signal. Suchlesser magnitude impulse noise masquerades as transmitted data. Pre-EQnoise-removal fails to detect (and hence fails to reduce) themasquerading impulse noise content. An aspect of the present inventionis that this problem can be overcome by reducing (if not removing)impulse noise after the received MCM signal has been equalized, in otherwords by performing a “post-E” type of impulse noise-removal. Post-Eimpulse noise-removal according to an aspect of the present inventioncan detect (and hence can reduce, if not remove) the masqueradingimpulse noise content.

For a received multiple carrier modulation (MCM) signal (a frequencydomain version of which is R) that corresponds to a transmitted MCMsignal (a frequency domain version of which is S), and has beenequalized (a “post-E” signal), embodiments of the present inventionreduce impulse-noise in the post-E signal. For example, the MCM signalcan be an orthogonal frequency-division multiplexing (OFDM) signal.Impulse noise in the post-E signal is estimated and then a portion ofthe impulse noise in the post-E signal is reduced (if not removed)according to the estimated impulse noise. The estimate of the impulsenoise is based upon an approximation of the total noise in the post-Esignal.

Other embodiments of the present invention remove the portion (ofimpulse noise in the post-E signal) also as a function of an estimatedchannel transfer function (Ĥ). For example, this can be done by takingthe matrix product (element-by-element product) of the estimated impulsenoise and an inverse (Ĥ⁻¹) of Ĥ, and then subtracting the matrix productfrom the equalized signal (the frequency domain version of which isR^((eq))).

FIG. 2 is a block diagram depicting an MCM, e.g., OFDM, system 200according to at least one embodiment of the present invention. Thesystem 200 includes a transmitter 202 and a receiver 224. Transmitter202 includes: a scrambling unit 204; a forward error correction (FEC)encoder 206; an interleaving unit 208; a quadrature amplitude modulation(QAM), phase-shift keying (PSK) (QAM/PSK) mapping unit 210; a pilotsignal insertion unit 212; an inverse fast Fourier transform (IFFT) unit214; and a guard-interval unit 216 to add a guard-interval. Where thecommunications channel is a free-space channel 222, transmitter 202 canfurther include a digital to analog (DAC) converter 218 and anup-converter 220; and the system 200 correspondingly includes antenna221 (associated with transmitter 202) and antenna 223 (associated withreceiver 224).

Receiver 224 includes: a guard-interval removing unit 230; a combinedFast Fourier Transform (FFT) & Equalization & Impulse-Noise-Compensationunit 250 (itself according to embodiments of the present invention); anoptional de-mapping unit 236; a de-interleaving unit 238; an FEC decoder240; and a de-scrambling unit 242. Where the communications channel isfree-space channel 222, receiver 224 further includes: a down-converter226; and an analog-to-digital converter (ADC) 228.

Typically, but not necessarily, the communication channel will befree-space channel 222; alternatively, the communications channel can bea wire, waveguide, etc. This is depicted in FIG. 2 by alternativecommunication channels 244 and 246. If communication channel 246 isused, then up-converter 220, antennas 221 and 223, and down-converter226 would not be present. If communication channel 244 is used, then DAC218 and ADC 228, as well as components 220-226, would no be present.

In MCM transmitter 202, a set of information bits passes throughscrambling unit 204, FEC encoder 206, interleaving unit 208, and then ismapped into baseband symbols {S_(k)} using modulation techniques such asphase-shift keying (PSK) and/or quadrature amplitude modulation (QAM) byQAM/PSK mapping unit 210. Pilot signal insertion is performed on theblock of baseband signals by pilot insertion unit 212. The output ofpilot insertion unit 212 is operated upon by IFFT unit 214. The outputof IFFT unit 214 is operated upon by guard interval unit 216 to addguard intervals, and the resultant signal is digital-to-analog convertedby DAC 218 (assuming the communications channel is either 222 or 246).

For each channel symbol interval, output of IFFT unit 214 can berepresented by the following time-domain equation. $\begin{matrix}{{s(t)} = {{\sum\limits_{k = 0}^{N - 1}{S_{k}{\mathbb{e}}^{{j2\pi}\quad k\quad\Delta\quad{ft}}\quad{where}\quad 0}} < t < T_{s}}} & (1)\end{matrix}$where:

-   -   N is the number of sub-carriers in the MCM signal;    -   Δf is the separation between adjacent sub-carriers; and    -   T_(s) is the channel symbol interval.

After passing through the communication channel, then beingdown-converted (assuming communication channel 222 is used) bydown-converter 226, then being digitized by ADC 228, then having theguard interval removed (by unit 230), and then being synchronized, theresultant signal (r_(k)) provided to unit 250 can be described by thefollowing discrete-time (as contrasted with continuous-time) equation.r _(k) =*h _(l) *s _(k) +n _(k) +u _(k)  (2)where:

-   -   k=0, 1, . . . , N−1;    -   s_(k)=s(kT_(s)/N).    -   h_(l) is the impulse response of the channel (l=1, . . . , L,        where L is the length of channel impulse response);    -   n_(k) is the additive white Gaussian noise (AWGN) term;    -   u_(k) is the impulse noise interference; and    -   “*” denotes convolution.

FIG. 3 is a more detailed block diagram of combinedFFT_&_Equalization_&_Impulse-Noise-Compensation unit 250 according to atleast one embodiment of the present invention. Unit 250 includes: a FFTunit 302; an equalizer unit 304; and an impulse-noise reducer unit 306.Impulse-noise reducer unit 306 includes: total-noise measuring unit 308;impulse-noise estimating unit 310; and compensated-signal generator 312.

FFT unit 302 operates upon the received MCM signal r_(k) that isobtained from guard unit 230 and outputs the frequency domain version R.The output R of FFT unit 302 is a series of data and pilot symbol valuesfor each symbol encoded on the received MCM signal r_(k). Relative toEquation No. 2, the output R can be represented in the frequency-domainby the following equationR=H·S+N+U  (3)where:

-   -   H is the discrete Fourier transform (DFT) of the channel's        impulse response;    -   S is the DFT of the transmitted MCM signal;    -   N is the DFT of the additive white Gaussian noise (AWGN) (also        known as background noise) term;    -   U is the DFr of the impulse noise term;    -   · denotes matrix (element-by-element) multiplication such that,        for S=(S0, S1, . . . ) and H=(H0, H1, . . . ), S·H={S₀H₀, S₁H₁,        . . . }; and    -   the bold style and UPPERCASE letters indicates VECTOR NOTATION        in the frequency domain.

It is assumed that the estimated channel transfer function (Ĥ) providedby equalizer 304 is approximately equal to the actual channel transferfunction (H), i.e. Ĥ≈H, and Ĥ⁻¹≈H⁻¹, and H⁻¹·H≈I, where I={1,1,1 . . .,1}.

Equalizer unit 304 of FIG. 3 operates upon R and outputs an equalizedversion R^((eq)) and an estimate Ĥ of the channel impulse response toimpulse-noise reducer unit 306. The following derivation yields theequation for the equalized signal R^(eq): $\begin{matrix}\begin{matrix}{R^{({eq})} = {R \cdot {\hat{H}}^{- 1}}} \\{= {\left( {{\hat{H} \cdot S} + N + U} \right) \cdot {\hat{H}}^{- 1}}} \\{= {{H \cdot S \cdot {\hat{H}}^{- 1}} + {\left( {N + U} \right) \cdot {\hat{H}}^{- 1}}}} \\{= {S + {\left( {N + U} \right) \cdot {\hat{H}}^{- 1}}}} \\{R^{({eq})} = {S + {D \cdot {\hat{H}}^{- 1}}}}\end{matrix} & (4)\end{matrix}$where

-   -   R^((eq)) is (again) the equalized version of the received signal    -   Ĥ⁻¹={Ĥ₀ ⁻¹, Ĥ₁ ⁻¹, . . . , ĤN-1 ⁻¹}; and    -   D=N+U is the total noise in the frequency domain.

Solving Eq. No. 4 for D and substituting Ĥ for H obtains the followingequation of estimated total noise, D.R ^((eq)) ·H=S·H+DR ^((eq)) ·H−S·H=D, assume H≈Ĥ{circumflex over (D)}=(R ^((eq)) −S)*Ĥ  (5)

Total-noise measuring unit 308 produces a frequency-domain estimate ofthe total noise ({circumflex over (D)}) and includes ademapping-and-pilot-insertion-unit 314; adder 316; and multiplier 318.Impulse-noise estimating unit 310 produces a frequency-domain estimateof impulse noise Û and includes: IFFT unit 322; a peaks detection unit326; and an FFT unit 328. Compensated-signal generator 312 performspost-E impulse-noise reduction (yielding frequency-domain signalR^((eq&comp))) and includes: an inversion unit 330; an optional delayunit 332; a multiplier 334; an optional delay unit 336; and an adder338.

In total-noise measuring unit 308, unit 314 receives equalized signalR^((eq)) and produces an estimated version Ŝ of the transmitted MCMsignal S, which gets provided to a negative input of adder 316. Adder316 also receives equalized signal R^((eq)). In other words, adder 316subtracts Ŝ from equalized signal R^((eq)) and outputs the difference.Multiplier 318 receives the difference R^((eq))−Ŝ and matrix-multipliesit with H to form an estimate of the total noise ({circumflex over(D)}), where {circumflex over (D)}=(R^((eq))−S)·Ĥ. Optional delay units,such as delay unit 332, are provided to accommodate delay introduced bythe Fast Fourier transformation, which is an example of delay introducedby signal processing. Such delay units are routinely added and/ormanipulated (as to operating characteristics) as part of implementingsignal processing technology in hardware according to the constraints ofa given set of circumstances.

In impulse-noise estimating unit 310, IFFT unit 322 receives total noiseestimate {circumflex over (D)} and outputs the time-domain version({circumflex over (d)}). Peaks-detection unit 326 operates upon thetotal noise estimate {circumflex over (d)} and generates a time-domainestimate of its impulse noise content (û). FFT unit 328 receives thetime-domain impulse-noise estimate û={û₀, û₁, . . . , û_(N-1)} andoutputs the frequency-domain version (Û).

The impulse noise can be estimated as follows (in the frequency-domain).$\begin{matrix}{{{\hat{U}}_{k} = {{{A_{t_{1}}{{\mathbb{e}}\left( {j\quad\frac{2\pi\quad{kt}_{1}}{N}} \right)}} + {A_{t_{2}}{{\mathbb{e}}\left( {j\quad\frac{2\pi\quad{kt}_{2}}{N}} \right)}} + \ldots + {A_{t_{N - 1}}{{\mathbb{e}}\left( {j\quad\frac{2\pi\quad{kt}_{L}}{N}} \right)}\quad{for}\quad k}} = 0}},1,{{\ldots\quad N} - 1}} & (6)\end{matrix}$where

-   -   L is the number of samples in one MCM symbol effected by        impulse-noise;    -   t₁, t₂, . . . , t_(L) are the positions of the samples; and    -   A₁, A₂, . . . , A_(L) are the complex amplitudes of the samples.

If both the total noise, D, and the AWGN, N, can be estimated({circumflex over (D)}, {circumflex over (N)}), then the impulse noise,U, can be estimated by solving the equation D=N+U as follows.Û={circumflex over (D)}−{circumflex over (N)}  (7)

But it is not easy to distinguish N from U in the frequency domain. SoEq. No. 7 is transformed back into the time-domain via inverse fastFourier transformation (IFFT). In terms of FIG. 3, IFFT unit 322 (again)operates upon total noise estimate {circumflex over (D)} output bytotal-noise measuring unit 308. Operation of peaks-detection unit 326will now be discussed.

To distinguish between impulse-noise samples u_(k) and AWGN-noisesamples n_(k), peaks-detection unit 326 can implement a rule ofdistinction. Such a rule uses an estimate of the variance, {circumflexover (σ)}, of {circumflex over (d)}, where {circumflex over (σ)} isgiven by the following equation. $\begin{matrix}{\hat{\sigma} = \sqrt{\frac{1}{N}{\sum\limits_{k = 0}^{N - 1}{d_{k}}^{2}}}} & (8)\end{matrix}$

An example of the distinction rule implemented by peaks-detection unit326 to distinguish between impulse noise samples u_(k) and AWGN noisesamples n_(k) is as follows. $\begin{matrix}{{\hat{u}}_{k} = \begin{Bmatrix}{{\hat{d}}_{k},} & {{{{for}\quad{{\hat{d}}_{k}}} > {C\quad\hat{\sigma}}}\quad} & {{k = 0},1,\ldots,{N - 1}} \\0 & {otherwise} & \quad\end{Bmatrix}} & (9)\end{matrix}$where

-   -   C is a threshold value corresponding to a small probability of        false detection.

FIG. 4 is a more detailed block diagram of peaks-detection unit 326according to at least one embodiment of the present invention.Peaks-detection unit 326 includes: a matrix-product-squared unit 402; aunit 404 to sum elements of a matrix; a scalar multiplier 406; acomparator 408; and a selective attenuator 410.

Matrix-product-squared unit 402 receives the time-domain version{circumflex over (d)}={{circumflex over (d)}_(k)} of the total noiseestimate {circumflex over (D)} from IFFT unit 322 and performs matrixmultiplication to take the square of {{circumflex over (d)}_(k)}, whichis {|{circumflex over (d)}_(k)|²}. Unit 404 operates upon the productfrom unit 402 to produce a scalar sum of the N elements in {|{circumflexover (d)}_(k)|²}, which is multiplied with $\frac{C^{2}}{N}$by multiplier 406 to produce product C{circumflex over (σ)}². Comparator408 compares each k^(th) member {|{circumflex over (d)}_(k)|²} againstthe product C{circumflex over (σ)}², and provides an indication eachcomparison to selective attenuator 410. Eq. No. 9 is implemented by unit410, which selectively sets û_(k)={circumflex over (d)}_(k) or û_(k)=0depending upon the corresponding k^(th) comparison in order to produce{û_(k)}=û.

In compensated-signal generator 312, inversion unit 330 receivesestimate Ĥ and outputs its inversion (Ĥ⁻¹). Delay unit 332 delaysinverted estimate Ĥ⁻¹. In some implementations, availability of Û mightbe delayed. Delay unit 332 can compensate for such delay.

Multiplier 334 receives the frequency-domain impulse-noise estimate Û(from FFT unit 338) and delayed inverted estimate Ĥ⁻¹ and multipliesthem together to form an intermediate product Û·Ĥ⁻¹, which is providedto a negative input of adder 338. Optional delay unit 336 (which wouldbe present if optional delay unit 332 is present) delays equalizedsignal R^((eq)). Adder 338 receives the delayed equalized signalR^((eq)) and, in effect, subtracts from it the intermediate productÛ·Ĥ⁻¹ to form the difference R^((eq&comp)), which is an equalized andimpulse-noise-reduced frequency-domain version of r_(k), and which isprovided to optional demapping unit 236, or optionally directly tode-interleaving 238.

Equalized and impulse-noise-reduced signal R^((eq&comp)) can be derivedas follows, where it is assumed that Û≈U. $\begin{matrix}\begin{matrix}{R^{({{{eq}\&}{comp}})} = {R^{({eq})} - {\hat{U} \cdot {\hat{H}}^{- 1}}}} \\{= {R^{({eq})} - {\left( {\hat{D} - \hat{N}} \right) \cdot {\hat{H}}^{- 1}}}} \\{= {R^{({eq})} - {\left( {{\left( {R^{({eq})} - \hat{S}} \right)\hat{H}} - \hat{N}} \right) \cdot {\hat{H}}^{- 1}}}} \\{= {R^{({eq})} - \left( {R^{({eq})} - \hat{S}} \right) + {\hat{N} \cdot {\hat{H}}^{- 1}}}} \\{= {\hat{S} + {\hat{N} \cdot {\hat{H}}^{- 1}}}} \\{R^{({{{eq}\&}{comp}})} = {\hat{S} + {\hat{N} \cdot {\hat{H}}^{- 1}}}}\end{matrix} & (10)\end{matrix}$

In FIG. 3, it is noted that equalizer unit 304 can estimate both Ĥ andĤ⁻¹. This lends itself to alternative arrangements ofFFT_&_Equalization_&_Impulse-Noise-Compensation unit 250, e.g., asfollows. In an alternative embodiment according to the presentinvention, inversion unit 330 is not provided; instead Ĥ⁻¹ can beprovided directly from equalizer unit 304 to delay unit 332 via signalpath 340. Another alternative embodiment according to the presentinvention has equalizer unit 304 providing Ĥ⁻¹ but not Ĥ, so an optionalinversion unit 342 is included. Inversion unit 342 receives Ĥ⁻¹ viasignal path 344 and provides Ĥ to multiplier 318.

FIG. 5 is a block diagram of receiver 224, at a level of detail betweenthat of FIGS. 2 and 3. As such, FIG. 5 can serve as a visual summary ofthe discussion of FIG. 3. In FIG. 5, equalizer 304 is shown as providingĤ, or Ĥ⁻¹ or Ĥ & Ĥ⁻¹, in keeping with the discussion above concerningalternative arrangements ofFFT_&_Equalization_&_Impulse-Noise-Compensation unit 250.

Though QAM/PSK mapping is performed by unit 210 of transmitter 202,demapping unit 236 is optional. If a hard-decision type of FEC decoder240 is used, then demapping unit 236 should be present. But if asoft-decision type of FEC decoder 240 is used, then demapping unit 236could be used but is not necessary. If interleaving unit 208 is notpresent in transmitter 202, then de-interleaving unit 238correspondingly would not be present in receiver 224. Similarly, ifscrambling unit 204 were not present in transmitter 202, thendescrambling unit 242 would not be present in receiver 224.

An optional clipping circuit 594 is depicted in FIG. 5, via phantomsignal paths, as being interposed between ADC 228 and unit 230,resulting in a variation 224′ of receiver 224, according to anotherembodiment of the present invention. In some circumstances, amplitudesof impulse-noise are so high that the aspect of demapping performed bydemapping-and-pilot-insertion unit 314 can become unreliable and theestimated impulse-noise sequence can differ significantly from the trueimpulse-noise sequence. This can be compensated by introducing pre-EQnoise-removal via clipping unit 594, which corresponds to clipping unit101 of the Background Art. While pre-EQ clipping itself introducesdistortion, it can be viewed as an additive impulse interference thatcan be compensated for byFFT_&_Equalization_&_Impulse-Noise-Compensation unit 250.

FIG. 6 is a block diagram of anFFT_&_Equalization_&_Impulse-Noise-Compensation unit 450 according toanother embodiment of the present invention. Unit 450 is similar to unit250, but adds one or more additional stages of impulse-noise reduction.The zeroith stage of impulse reduction corresponds to units 302, 304 and406 ₀, where impulse-noise reducer unit 406 ₀ corresponds toimpulse-noise reducer unit 306 and produces the zeroith iteration of theequalized and impulse-noise-reduced signal, namely R₀ ^((eq&comp)). Thefirst stage of impulse-noise reduction corresponds to impulse-noisereducer unit 406 ₁, which operates upon R₀ ^((eq&comp)) and produces thefirst iteration of the equalized and impulse-noise-reduced signal,namely R₁ ^((eq&comp)). Unit 450 includes, as part of the first stage,an optional delay unit 452 ₀ that correspondingly delays Ĥ and/or Ĥ⁻¹according to delay induced by the processing performed by unit 406 ₀.Similarly, unit 450 includes, as part of the second stage, an optionaldelay unit 452 ₁.

FIG. 6 depicts a total of P stages, the last stage being stage P-1 thatincludes impulse-noise reducer unit 406 _(P-1). It is noted that stageP-1 does not include a delay unit as it is the final stage ofimpulse-noise reduction. In other words, in FIG. 6, there are P stagesof impulse-noise reduction but P-1 delay units delay unit 452 ₁. Anadvantage of multi-stage unit 450 is that impulse-noise estimation isnot perfect, but iteratively estimating impulse-noise andcorrespondingly compensating cumulatively will achieve better noisereduction than a single stage of impulse-noise estimation andcompensation. As a practical matter, the choice of single-stage versusmultiple-stages of impulse-noise reduction depends upon thecircumstances of the use to which anFFT_&_Equalization_&_Impulse-Noise-Compensation unit is applied.

FIG. 7A is a block diagram of an alternative toFFT_&_Equalization_&_Impulse-Noise-Compensation unit 250, namelyFFT_&_Equalizationr_&_Impulse-Noise-Compensation unit 750, according toanother embodiment of the present invention. Similarities between unit750 and unit 250 are reflected in the reuse of the same item numbers orcorresponding item numbers, e.g., 310⇄710. Unit 750 includes: Amongother differences, unit 250 includes FFT units 302 and 328 and IFFR unit322, while unit 750 does not include an IFFT unit but instead variouslyuses one FFT unit for both FFT and IFFT.

Unit 750 includes an FFT_&impulse-noise-reducer unit 706 and anequalizer unit 304. FFT_&_impulse-noise-reducer unit 706 includes:FFT-Reuse_&_Impulse-Estimating unit 710; total-noise measuring unit 308;and compensated-signal generator 312. FFT-Reuse_&_Impulse-Estimatingunit 710 includes: a multiplexer (mux) 760; FFT unit 762; ademultiplexer (demux) 764; matrix complex conjugate units 766 and 768; ascalar source 770; a multiplier 772; and peaks-detection unit 326.

Mux 760 and demux 764 are controlled to select the same input. During afirst phase in operation of unit 710, mux 760 and demux 764 arecontrolled to select signals at the first input, consequently signalr_(k) is received from guard unit 230 and operated upon by FFT unit 762,which outputs frequency domain version R. Total noise estimate{circumflex over (D)} is received by conjugate unit 766, which producesthe transpose {circumflex over (D)}* and provides it to the second inputof mux 760.

During a second phase in operation of unit 710, mux 760 and demux 764are controlled to select signals at the second input, consequently FFTunit 762 operates upon {circumflex over (D)}* and producesFFT({circumflex over (D)}*). Conjugate unit 768 operates upon thesignals on the second output of demux 764, which produces the signal[FFT({circumflex over (D)}*)]. Scalar source, e.g., a location inmemory, provides the scalar value $\frac{1}{N},$which multiplier 772 multiplies together with signal [FFT({circumflexover (D)}*)]^(*) to form time-domain total-noise estimate {circumflexover (d)} as the product. In other words, units 768-772 operate togetherto perform an IFFT according to the following equation. $\begin{matrix}{{{IFFT}(x)} = {\frac{1}{N}\left\lbrack {{FFT}\left( x^{*} \right)} \right\rbrack}^{*}} & (11)\end{matrix}$As in unit 250, peaks-detection unit 326 operates upon d to producetime-domain impulse-noise estimate û.

During a third phase in operation of unit 710, mux 760 and demux 764 arecontrolled to select signals at the third input, consequently FFT unit762 operates upon u and produces frequency-domain impulse-noise estimateÛ. The third output of demux 764 provides Û to multiplier 334 incompensated-signal generator 312.

The three phases of operation of unit 710 should be completed before thenext MCM symbol, e.g., OFDM symbol, appears at the first input of mux760. Accordingly, a clock provided to FFT unit 762 as used in unit 710can be made three times faster than the clock provided to FFT unit 302as used in unit 250. Alternatively, delay associated with the threephases of operation of unit 710 can be compensated for by delay units332 and 336.

In FIG. 7A, as in FIG. 3, it is noted that equalizer unit 304 canestimate both Ĥ and Ĥ⁻¹. This lends itself to alternative arrangementsof FFT_&_Equalization_&_Impulse-Noise-Compensation unit 750 thatcorrespond to the alternative arrangements of unit 250 depicted in FIG.3. Such alternative arrangements are not depicted in FIG. 7A, forsimplicity of illustration. FIG. 7B is a block diagram visual-summary ofa receiver 780 that corresponds to receiver 224 of FIG. 5. It is notedthat components 226-230 and 236-242 of transmitter 224 are included intransmitter 780 but are not explicitly depicted in FIG. 7B, forsimplicity of illustration; inclusion of components 226-230 and 236-242is indicated by the ellipses ( . . . ). As with the depiction ofequalizer 304 in FIG. 5, equalizer 304 in FIG. 7B is shown as providingĤ, or Ĥ⁻¹ or Ĥ & Ĥ⁻¹, in keeping with the discussion above concerningalternative arrangements ofFFT_&_Equalization_&_Impulse-Noise-Compensation unit 250.

FIG. 8A is a block diagram of aFFT_&_Equalization_&_Impulse-Noise-Compensation unit 850 according to atleast one embodiment of the present invention. Unit 850 is similar tounit 450 in having multiple stages but exhibits with FFT-reuse as inunit 750.

Compensated signal generator 812 corresponds to compensated signalgenerator 312 but includes a multiplexer (mux) 882 interposed betweenequalizer unit 304 and delay unit 336. The first input of mux 882 isconnected to equalizer unit 304, while the second input is feedbackconnected to the output of adder 338 via an optional delay unit 884.

The first phase of the zeroith stage of unit 806 is represented by mux882 being controlled to select its first input, thereby passing R^((eq))to delay unit 336, etc. As such, the first phase of the zeroith stage ofunit 806 corresponds to the first phase in operation of unit 710. Thesecond and third phases of the zeroith stage of unit 806 corresponds tothe second and third phases in operation of unit 710.

After the zeroith stage of unit 806, mux 882 is connected to the outputof delay unit 884. The connection of mux 882 to delay unit 884 ismaintained in the stages of unit 806 that follow. A clock provided toFFT unit 762 (within unit 710) is also used in unit 850 but should bemade 2M+1 times faster than the clock provided to FFT unit 302 used inunit 250.

Also, mux 890 and an optional delay unit 892 are included for delay of Ĥduring iteration operations similar to the inclusion of delay units 452i in FIG. 6. In the zeroith stage, mux 890 connects to equalizer 304. Instages that follow, mux 890 connects to delay unit 892.

Alternatively, because the equalizer 304 itself may have memory to storechannel estimation, mux 890 and delay unit 892 may be implemented insideequalizer 304.

Similarly, the zeroith stage of impulse reduction via unit 850 operatesupon R^((eq)) and produces the zeroith iteration of the equalized andimpulse-noise-reduced signal, namely R₀ ^((eq&comp)). The first stage ofimpulse-noise reduction operates upon R₀ ^((eq&comp)) and produces thefirst iteration of the equalized and impulse-noise-reduced signal,namely R₀ ^((eq&comp)); etc. Again, an advantage of multi-stage unit 450is that impulse-noise estimation is not perfect, but iterativelyestimating impulse-noise and correspondingly compensating cumulativelywill achieve better noise reduction than a single stage of impulse-noiseestimation and compensation. As a practical matter, the choice ofsingle-stage versus multiple-stages of impulse-noise reduction dependsupon the circumstances of the use to which anFFT_&_Equalization_&_Impulse-Noise-Compensation unit is applied.

FIG. 8B is a block diagram visual-summary of a receiver 880 thatcorresponds to receiver 780 of FIG. 7B. As with the depiction ofequalizer 304 in FIG. 7B, equalizer 304 in FIG. 8B is shown as providingĤ, or Ĥ⁻¹ or Ĥ & Ĥ⁻¹. Also, a feedback path 886 represents the second,third, etc. stages of unit 850.

Impulse-noise compensation can also take place in the time-domain viaFFT_&_Equalization_&_Impulse-Noise-Compensation unit 950 of FIG. 9A,according to another embodiment of the present invention. Unit 950 ishas similarities to FFT_&_Equalization_&_Impulse-Noise-Compensation unit250, as reflected in the reuse of the same item numbers or correspondingitem numbers, e.g., 308⇄908. Unit 950 includes: FFT unit 302; equalizerunit 304; and an impulse-noise reducer unit 906. Impulse-noise reducerunit 906 includes: total-noise measuring unit 908; impulse-noiseestimating unit 911; and compensated signal generator 913.

Total-noise measuring unit 908 produces a time-domain estimate of thetotal noise ({circumflex over (d)}) and includes a demapping andpilot-insertion unit 314; a multiplier 919; IFFT unit 332; and an adder917. Impulse-noise estimating unit 911 produces a time-domain estimateof impulse noise content û (and includes: peaks-detection unit 326.Compensated-signal generator 913 produces frequency-domain equalized andcompensated signal R^((eq&comp)) and includes: an adder 939; an FFT unit929; optional delay unit 330; inversion unit 332; and a multiplier 935.

Recalling Eq. No. 2 and d_(k)=n_(k)−u_(k), then solving for n_(k),time-domain total-noise estimate d can be derived as follows.$\begin{matrix}\left. \begin{matrix}{r_{k} = {{h_{l}*s_{k}} + \left( {d_{k} - u_{k}} \right) + u_{k}}} \\{= {{h_{l}*s_{k}} + d_{k}}}\end{matrix}\quad\Downarrow\begin{matrix}{{\hat{d}}_{k} = {r_{k} - \left( {{\hat{s}}_{k}*{\hat{h}}_{k}} \right)}} \\{{= {r_{k} - {{IFFT}\left( {\hat{S} \cdot \hat{H}} \right)}}},{{{where}\quad k} = 0},1,\ldots\quad,{N - 1}}\end{matrix} \right. & (12)\end{matrix}$

In total-noise measuring unit 908, unit 314 receives equalized signalR^((eq)) and produces an estimated version Ŝ of the transmitted MCMsignal S, which gets provided to multiplier 919, which multiplies ittogether with Ĥ to form frequency domain matrix product Ŝ·Ĥ. Thatproduct is transformed by IFFT unit 332 into the time-domain versionŝ*ĥ, where “*” denotes convolution and provided to a negative input ofadder 917. Adder 917 also receives, via optional delay unit 937, adelayed version of r_(k). In other words, adder 917 subtracts ŝ*ĥ fromequalized signal r_(k) and outputs the difference, which is time-domainestimate of the total-noise {circumflex over (d)}. If IFFT unit 332introduces delay, then delay unit 937 is provided, and correspondinglyso is delay unit 330.

Peaks-detection unit 326 of impulse-noise estimating unit 911 receives{circumflex over (d)} and generates the time-domain estimate of itsimpulse noise content û.

A negative input of adder 939 of compensated signal generator 913receives the time-domain estimate of impulse noise content û. Adder 939also receives, via delay unit 937, the delayed version of r_(k). Inother words, adder 939 subtracts û from equalized signal r_(k) andoutputs the difference, which is time-domain compensated signalr^((comp)). FFT unit 929 receives r^((comp)) and produces thefrequency-domain version R^((comp)). Optional delay unit 330 receivesand delays Ĥ. Inversion unit 332 operates upon the delayed version of Ĥand outputs delayed (Ĥ⁻¹). Multiplier 935 receives R^((comp)) andmatrix-multiplies it with delayed Ĥ⁻¹ to form frequency-domain equalizedand compensated signal R^((eq&comp)).

In FIG. 9A, as in FIG. 3, it is noted that equalizer unit 304 canestimate both Ĥ and Ĥ⁻¹. This lends itself to alternative arrangementsof FFT_&_Equalization_&_Impulse-Noise-Compensation unit 950 thatcorrespond to the alternative arrangements of unit 250 depicted in FIG.3. Such alternative arrangements are not depicted in FIG. 9A, forsimplicity of illustration. FIG. 9B is a block diagram visual-summary ofa receiver 980 that corresponds to receiver 224 of FIG. 5. It is notedthat components 226-230 and 236-242 of receiver 224 are included intransmitter 980 but are not explicitly depicted in FIG. 9B, forsimplicity of illustration; inclusion of components 226-230 and 236-242is indicated by the ellipses ( . . . ). As with the depiction ofequalizer 304 in FIG. 5, equalizer 304 in FIG. 9B is shown as providingĤ, or Ĥ⁻¹ or Ĥ & Ĥ⁻¹. Also, a feed-forward path 988 represents receivedMCM signal r_(k) being directly input to unit 950 (via optional delayunit 937, etc.).

FIG. 10 is a block diagram of aFFT_&_Equalization_&_Impulse-Noise-Compensation unit 1050 according toat least one embodiment of the present invention. Unit 1050 is similarto unit 450 in terms of being multi-staged. The zeroith stage of impulsereduction corresponds to units 302, 304 and 1006 ₀, where impulse-noisereducer unit 1006 ₀ produces the zeroith iteration of the equalized andimpulse-noise-reduced signal, namely R₀ ^((eq&comp)). The first stage ofimpulse-noise reduction corresponds to impulse-noise reducer unit 1006₁, which operates upon R₀ ^((eq&comp)) and produces the first iterationof the equalized and impulse-noise-reduced signal, namely R₁^((eq&comp)). Unit 1050 includes, as part of the first stage, a delayunit 452 ₀ that correspondingly delays Ĥ and/or Ĥ⁻¹ and an optionaldelay unit 1054 ₀ that correspondingly delays r_(k), according to delayinduced by the processing performed by unit 1006 ₀. Similarly, unit 450includes, as part of the second stage, delay unit 452 ₁ and optionaldelay unit 1054 ₁.

FIG. 10 depicts a total of P stages, the last stage being stage P-1 thatincludes impulse-noise reducer unit 1006 _(P-1). It is noted that stageP-1 does not include delay units as it is the final stage ofimpulse-noise reduction. In other words, in FIG. 6, there are P stagesof impulse-noise reduction but P-1 delay units 452, and 1054 ₁. Anadvantage of multi-stage unit 1050 is that impulse-noise estimation isnot perfect, but iteratively estimating impulse-noise andcorrespondingly compensating cumulatively will achieve better noisereduction than a single stage of impulse-noise estimation andcompensation. Again, as a practical matter, the choice of single-stageversus multiple-stages of impulse-noise reduction depends upon thecircumstances of the use to which anFFT_&_Equalization_&_Impulse-Noise-Compensation unit is applied.

FIG. 11A is a block diagram of an alternativeFFT_&_Equalization_&_Impulse-Noise-Compensation unit (relative to unit950), according to at least one embodiment of the present invention. TheFFT unit in FIG. 11A is re-used similarly to how FFT unit 762 is re-usedin FIG. 7A.

FIG. 11B is a block diagram visual-summary of a receiver 1180, relatedto FIG. 11A, according to at least one embodiment of the presentinvention.

FIG. 12A is a block diagram of an alternativeFFT_&_Equalization_&_Impulse-Noise-Compensation unit (relative to thecorresponding unit in FIG. 11A), according to at least one embodiment ofthe present invention. The FFT unit in FIG. 12A is re-used similarly tohow the FFT unit of FIG. 11A is re-used. Relative to FIG. 11A, muxes1282, 1292 and 1294 are included in FIG. 12A, which permits theFFT_&_Equalization_&_Impulse-Noise-Compensation unit of FIG. 12A toperform iterations in time in a manner corresponding to unit 1050 ofFIG. 10.

FIG. 12B is a block diagram visual-summary of a receiver 1280, relatedto FIG. 12A, according to at least one embodiment of the presentinvention.

The operation of FIG. 11A and FIG. 12A similarly, respectively, similarto that of FIG. 9A, FIG. 10, FIG. 7A and FIG. 8A, thus detailedexplanations are omitted herein for brevity.

It is noted that each of the receivers, e.g., 780, 880, 980, 1180 and1280 can also be modified to include optional clipping circuit 594resulting in receiver variations 780′, 880′, 980′, 1180′ and 1280′,according to other embodiments of the present invention.

Embodiments of the present invention perform better than the BackgroundArt. For example, embodiments of the present invention can enjoy up toabout 5 dB relative improvement in symbol error rate percentage for agiven symbol as compared to the Background Art. Also, the relativeimprovement is inversely (and approximately linearly) proportional to aratio of signal to total noise.

The present invention being thus described, it will be obvious that thesame may be varied in many ways. Such variations are not to be regardedas a departure from the spirit and scope of the present invention, andall such modifications are intended to be included within the scope ofthe present invention.

1. A method of reducing noise in a multiple carrier modulated (MCM)signal that has been equalized, the method comprising: estimatingimpulse noise based in the equalized signal; and removing a portion ofthe noise upon the equalized signal as a function of the estimatedimpulse noise.
 2. The method of claim 1, wherein themulti-carrier-modulated signal is an orthogonal frequency-divisionmultiplexing (OFDM) signal.
 3. The method of claim 1, wherein theremoving step removes the portion also as a function of an estimatedchannel transfer function (Ĥ).
 4. The method of claim 3, wherein atleast part of the removing step takes place in a frequency domain. 5.The method of claim 4, wherein the removing step removes the portion bytaking the matrix product of the estimated impulse noise and an inverse(Ĥ⁻¹) of Ĥ, and subtracting the product from the equalized signal. 6.The method of claim 3, wherein at least a part of the removing steptakes place in a time domain.
 7. The method of claim 6, wherein theremoving step includes subtracting the time-domain approximated impulsenoise from the received signal to form a compensated version of thereceived-signal.
 8. The method of claim 7, wherein the removing stepfurther includes taking the fast Fourier transform (FFT) of thetime-domain compensated received-signal to produce a frequency-domainversion of the compensated received-signal, and taking the product ofthe frequency-domain version of the compensated received-signal and aninverse (Ĥ⁻¹) of Ĥ.
 9. The method of claim 1, wherein the estimatingstep includes approximating total noise in the equalized signal, andapproximating the impulse noise based upon the approximated total noise.10. The method of claim 9, wherein at least part of the step ofapproximating the impulse noise takes place in a time domain.
 11. Themethod of claim 10, wherein the step of approximating the impulse noiseincludes: using peak-detection to produce a time-domain version of theestimated impulse noise based upon a time-domain version of theapproximated total noise.
 12. The method of claim 9, wherein at leastpart of the step of approximating the total noise takes place in afrequency domain.
 13. The method of claim 12, wherein the step ofapproximating the total noise includes: estimating a baseband signalthat includes a set of transmitted symbols; subtracting the estimatedbaseband signal from the equalized signal to form a set of differences;and multiplying the set of differences by an estimated channel transferfunction (Ĥ).
 14. The method of claim 9, wherein at least part of thestep of approximating the total noise takes place in a time domain. 15.The method of claim 14, wherein the step of approximating the totalnoise includes: estimating a baseband signal that includes a set oftransmitted symbols; taking the matrix product of the baseband signaland an estimated channel transfer function (Ĥ) to form afrequency-domain product; taking the inverse fast Fourier transform(IFFT) of the frequency-domain product to form a time-domain version ofthe product; subtracting the time domain product from the receivedsignal to form a time-domain version of the estimated total noise. 16.The method of claim 1, wherein: the estimating step and the removingstep can be performed iteratively, a first such iteration resulting in afirst noise-reduced version of the equalized signal; and the methodfurther including making a second iteration of the estimating step andthe removing step in which the estimating step operates upon the firstnoise-reduced version of the equalized signal; the second iterationproducing a second noise-reduced version of the equalized signal whichhas a lower noise content than the first version.
 17. The method ofclaim 16, further comprising: making a third iteration of the estimatingstep and the removing step in which the estimating step operates uponthe second noise-reduced version of the equalized signal; wherein thethird iteration produces a third noise-reduced version of the equalizedsignal which has a lower noise content than the second version.
 18. Themethod of claim 1, further comprising: clipping, prior to equalizing theMCM signal, peaks above a threshold; wherein the equalized signal is anequalized version of the clipped MCM signal.
 19. The method of claim 18,wherein the clipping step clips the MCM signal to either a thresholdlevel or to zero.
 20. An apparatus for reducing noise in a receivedmultiple carrier modulated (MCM) signal, the apparatus comprising: aFourier transformer operable upon the received MCM signal; an equalizeroperable to equalize a Fourier-transformed signal from the Fouriertransformer; and a total-noise estimator operable to estimate a totalnoise in the equalized signal from the equalizer; an impulse-noiseestimator operable to estimate impulse noise based upon the estimatedtotal-noise; and a noise compensator operable to remove a portion ofimpulse-noise on the equalized signal as a function of the estimatedimpulse-noise.
 21. The apparatus of claim 20, wherein the MCM signal isan orthogonal frequency-division multiplexing (OFDM) signal.
 22. Theapparatus of claim 20, wherein the noise compensator is operable also asa function of an estimated channel transfer function (Ĥ).
 23. Theapparatus of claim 22, wherein removal by the noise compensator is in afrequency domain.
 24. The apparatus of claim 23, wherein the noisecompensator is operable to remove by taking the matrix product of theestimated impulse noise and an inverse (Ĥ⁻¹) of Ĥ, and subtracting theproduct from the equalized signal.
 25. The apparatus of claim 22,wherein removal by the noise compensator is in a time domain.
 26. Theapparatus of claim 25, wherein the noise compensator is further operableto remove by subtracting the time-domain approximated impulse noise fromthe received MCM signal in the time domain to form a compensated signal.27. The apparatus of claim 26, wherein the noise compensator is furtheroperable to: take the fast Fourier transform (FFT) of the time-domaincompensated signal to produce a frequency-domain version of thecompensated signal; and take the product of the frequency-domain versionof the compensated signal and an inverse (Ĥ⁻¹) of Ĥ.
 28. The apparatusof claim 20, wherein the impulse-noise estimator is operable to estimatethe impulse noise in the time domain.
 29. The apparatus of claim 28,wherein the impulse-noise estimator is operable to estimate by usingpeak-detection to produce a time-domain version of the estimated impulsenoise based upon a time-domain version of the approximated total noise.30. The apparatus of claim 20, wherein the total-noise estimator isoperable to provide the estimated total noise in the frequency domain.31. The apparatus of claim 30, wherein the total-noise estimator isoperable to approximate the total noise by: estimating a baseband signalthat includes a set of transmitted symbols; subtracting the estimatedbaseband signal from the equalized signal to form a set of differences;and multiplying the set of differences by an estimated channel transferfunction (Ĥ), respectively.
 32. The apparatus of claim 20, wherein thetotal-noise estimator is operable to provide the estimated total noisein the time domain.
 33. The apparatus of claim 32, wherein thetotal-noise estimator is operable to approximate the total noise by:estimating a baseband signal that includes a set of transmitted symbols;taking the matrix product of the baseband signal and an estimatedchannel transfer function (Ĥ) to form a product; taking the inverse fastFourier transform (IFFT) of the product to form a time-domain version ofthe product; subtracting the time domain product from the receivedsignal to form a time-domain version of the estimated total noise. 34.The apparatus of claim 20, wherein one of the following applies: theequalizer is operable to determine an inverse (Ĥ⁻¹) of an estimatedchannel transfer function (Ĥ) and the noise compensator is operable toinvert Ĥ⁻¹ to produce Ĥ; the equalizer is operable to determine Ĥ andthe noise compensator is operable to produce Ĥ⁻¹; and the equalizer isoperable to produce both Ĥ⁻¹ and Ĥ.
 35. The apparatus of claim 34,wherein: the total-noise estimator, the impulse-noise estimator and thenoise compensator are arranged in a first stage and the noise-reducedversion of the equalized signal is a first such version; and theapparatus further includes at least a second stage having correspondinga second total-noise estimator operable upon the first noise-reducedversion of the equalized signal fed back thereto, a second impulse-noiseestimator, and a second noise compensator operable to output a secondnoise-reduced version of the equalized signal which has a lower noisecontent than the first version.
 36. The apparatus of claim 35, whereinthe second total-noise estimator is also operable upon the receivedsignal fed forward thereto.
 37. The apparatus of claim 35, wherein theapparatus further comprises at least a third stage having acorresponding third total-noise estimator operable upon the secondnoise-reduced version of the equalized signal fed back thereto, a thirdimpulse-noise estimator and a third noise compensator operable to outputa third noise-reduced version of the equalized signal which has a lowernoise content than the second version.
 38. The apparatus of claim 37,wherein the second total-noise estimator is also operable upon thereceived signal fed forward thereto.
 39. The apparatus of claim 20,wherein: the apparatus further comprises a first fast Fouriertransformer (FFT) to provide a frequency-domain version of the receivedsignal to the equalizer; and the impulse-noise estimator includes aninverse FFT (IFFT) and a second FFT, the IFFT providing a time-domainversion of the total noise, the impulse-noise estimator being operableto provide a time-domain estimate of the impulse noise based upon thetime-domain estimated total noise, and the second FFT being operable toprovide a frequency-domain version of the estimated impulse noise. 40.The apparatus of claim 20, wherein: the impulse noise estimator isoperable, in part, to make an inverse fast Fourier (IFF) transformation;the noise compensator is operable, in part, to make a fast Fourier (FF)transformation; the apparatus further comprises a fast Fouriertransformer (FFT); the apparatus being configured to selectively connectthe FFT according to at least three layouts, the first layout havingconnections such that operation of the FFT can provide afrequency-domain version of the received signal to the equalizer, thesecond layout having connections such that operation of the FFT can forma part of the IFF transformation, and the third layout havingconnections such that operation of the FFT can form a part of the FFtransformation.
 41. The apparatus of claim 40, wherein: the first,second and third layouts are part of a first arrangement and thenoise-reduced version of the equalized signal is a first such version;and the apparatus further being organized to selectively adopt a atleast a second arrangement in which the second layout operates upon thefirst noise-reduced version of the equalized signal fed back thereto;and the noise compensator in the second arrangement is operable tooutput a second noise-reduced version of the equalized signal which hasa lower noise content than the first version.
 42. The apparatus of claim41, wherein: the apparatus is further being organized to selectivelyadopt at least a third arrangement in which the second layout operatesupon the second noise-reduced version of the equalized signal fed backthereto; and the noise compensator in the third arrangement is operableto output a third noise-reduced version of the equalized signal whichhas a lower noise content than the second version.
 43. An apparatus forreducing noise in a multi-carrier-modulated (MCM) signal, the apparatuscomprising: a down-converter; an analog to digital converter to digitizethe output of the down-converter; a guard-interval removing unitoperable upon the digitized output of the down-converter; and a combinedFFT, equalization and impulse-noise-compensation unit operable upon asignal from the guard-interval-removing unit.
 44. The apparatus of claim43, wherein the combined FFT, equalization andimpulse-noise-compensation unit includes: an equalizer operable uponsignal from the guard-interval removing unit; a total-noise estimatoroperable upon a signal from the equalizer; an impulse-noise estimatoroperable upon a signal from the total-noise estimator; and a noisecompensator operable upon the signal from the equalizer and the signalfrom the impulse-noise estimator.
 45. The apparatus of claim 43, whereinthe multi-carrier-modulated signal is an orthogonal frequency-divisionmultiplexing (OFDM) signal.
 46. A method of reducing noise in a receivedmultiple carrier modulated (MCM) signal that has been partiallyequalized, the method comprising: estimating impulse noise based uponthe partially-equalized signal; and removing a portion of the noise inthe received signal in the time-domain as a function of the estimatedimpulse noise.
 47. The method of claim 46, wherein: the removing stepproduces a time-domain compensated signal; and the method furthercomprises equalizing a frequency-domain version of the compensatedsignal.
 48. The method of claim 47, wherein the equalizg step equalizesas a function of an estimated channel transfer function (Ĥ).